This post authored by John Coonrod, Technical Marketing Manager, and team originally appeared on the ROG Blog hosted by Microwave Journal.

Millimeter-wave frequencies were once few and far-between, in terms of applications and circuits using frequencies above 30 GHz. But that is about to change quickly, with Fifth Generation (5G) wireless networks and automotive radar systems both incorporating millimeter-wave frequency bands. For many circuit designers, these frequencies may represent uncharted territory and may require some thought not only about a suitable printed-circuit-board (PCB) material, but of the optimum transmission-line technology, board layouts, and connector launches. Many circuit designers face new challenges with the inevitable increase of millimeter-wave applications.

Circuit designers familiar with a particular transmission-line technology may ask: Can’t I stick with microstrip at these higher frequencies, if the PCB material delivers the performance I need? Microstrip is widely used in circuits from about 300 MHz to 30 GHz. Above 30 GHz, at millimeter-wave frequencies (30 to 300 GHz), microstrip suffers increased radiation loss and problems with spurious propagation modes. Designers working on circuits with both microwave and millimeter-wave transmission lines will often make a transition from microstrip to grounded coplanar-waveguide (GCPW) transmission lines which, when designed and fabricated properly, have little or no radiation loss and minimal spurious mode propagation.

For circuits with wideband coverage and without transitions between different transmission-line technologies, stripline is often used from lower microwave frequencies to millimeter-wave frequencies. However, forming a signal launch from a coaxial connector to stripline on a PCB has never been easy at microwave frequencies, and can become more challenging at higher, millimeter-wave frequencies with the shrinking dimensions of transmission-line structures. Ideally, the transition from the coaxial domain of a high-frequency connector to the parallel plane of a stripline PCB should be smooth, with little or no signal loss or reflections and no spurious modes. Assuming well-matched signal launches, stripline can be an excellent choice of  transmission line for millimeter-wave PCBs, although circuit fabrication is somewhat more involved than when forming microstrip or GCPW transmission lines.

Easier to Build?

Microstrip and GCPW circuits are attractive for their ease of assembly, each with a single dielectric layer with ground plane on bottom and signal conductors and components on top. Since the circuitry is exposed, components can be attached directly to the transmission lines on the signal plane. Stripline, on the other hand, surrounds its signal conductors with dielectric layers which in turn have ground planes on top and bottom. Because stripline’s signal conductors are buried in a multiple-layer circuit assembly, making connections between components and the signal conductors is never routine. Signal connections in microwave stripline PCBs are typically made by means of conductive viaholes: holes drilled through the dielectric layers and plated with conductive metal. Plated viaholes, or plated through holes (PTHs) as they are known, provide short, electrically conductive signal paths through the dielectric layers but also add their own capacitance and inductance values to the circuitry, impacting performance at higher frequencies. They become part of a circuit diagram (which must be modeled) at millimeter-wave frequencies.

Effective use of stripline transmission-line technology for millimeter-wave PCBs depends on finding the optimum plated viahole structure for low-loss, low-reflection transmission of high-frequency signals to the embedded signal plane. The transition provided by well-formed viaholes through stripline circuitry is essential not only for energy from signal-launch connectors but any electrical connections made to and from external components.

Laser technology can be an effective means of forming the small viaholes, or microvias, needed for stripline PCB interconnections at millimeter-wave frequencies. Precisely controlled laser drilling systems are designed to cut micron-sized microvias by burning through the top copper ground plane of a stripline circuit assembly, through the dielectric material beneath it, and to the signal plane lying between the two dielectric layers. Copper plating is applied and, in this way, a conductive path is formed through the hole from the top copper layer to the signal plane beneath. Such microvias can be formed with extremely small diameters and with the short lengths needed for thin dielectric materials typically used at smaller-wavelength, millimeter-wave frequencies.

By using this commercially available laser-based microvia-forming process, excellent performance can be achieved in stripline interconnections at millimeter-wave frequencies. Larger PTHs formed in stripline circuit assemblies can add unwanted capacitances and inductances at millimeter wavelengths, even in the shortest lengths.

Low-loss, low-reflection signal launches in stripline have been commonly realized in circuits for use to about 40 GHz; it can be difficult to achieve the good match and construction between connector interface and viahole for stripline circuits with coaxial launches at frequencies above 40 GHz. However, the choice of PCB material can play a role in the effectiveness of stripline circuits at millimeter-wave frequencies, based on recent experiments with RO3003™ laminates from Rogers Corp. Using these materials with standard stripline transmission-line structures, low-loss coaxial signal launches were measured to as high as 60 GHz. With several minor modifications, it should be possible to achieve practical coaxial-to-stripline signal launches out to 80 GHz using these same circuit materials.

When considering a PCB material that can support microvias for millimeter-wave circuits, stability at those higher frequencies is a key requirement. RO3003 circuit material has shown excellent mechanical and electrical stability above 30 GHz. It is mechanically stable, with the stability typically realized on other glass-reinforced materials as part of a multilayer construction. However, RO3003 laminates do not use glass reinforcement, so microvias can be laser-formed reliably and consistently without the effects from the lasered glass. RO3003 features coefficient of thermal expansion (CTE) closely matched to that of copper, so that microvias remain structurally and electrically sound even with thermal cycling. Regardless of the choice of transmission line, RO3003 circuit material, with its consistent dielectric constant (Dk) and dissipation factor (Df) over a wide range of frequencies, is a logical starting place for those higher-frequency circuits.

While there may not be one perfect transmission-line technology for millimeter-wave circuits, the choice of a starting point—the PCB material—can make a difference in the final performance possible at those higher frequencies. Microstrip and GCPW technologies support many millimeter-wave circuit applications with ease of fabrication and testing, but it has been shown that stripline is capable of excellent circuit performance at millimeter-wave frequencies when teamed with the right circuit materials. 

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This post authored by John Coonrod originally appeared on the ROG Blog hosted by Microwave Journal.

Bandpass filters are essential to many RF/microwave circuits and systems. They eliminate unwanted signals and noise, and can work with both receivers and transmitters. Bandpass filters can be assembled in a variety of ways, using lumped-element discrete inductors and capacitors at lower frequencies and semiconductor technologies for tiny monolithic filters at higher frequencies. Still, the most popular RF/microwave bandpass filters may be the ones based on microstrip transmission lines with distributed circuit elements on printed-circuit-board (PCB) substrates. With the right circuit material, microstrip bandpass filters can provide excellent performance in small circuits. This first of two blogs on RF/microwave bandpass filters will review some of their basic performance parameters and how they relate to PCB material characteristics, with a focus on one material in particular, RT/duroid® 6010.2LM circuit material from Rogers Corp. As a followup, the next blog will explore how bandpass filters perform on other circuit materials.

A bandpass filter is defined by a center frequency within a passband, channeling all signals within that passband with minimal loss while rejecting signals at frequencies above and below the passband with as much attenuation as possible. In contrast, a lowpass filter passes all signals below a given cutoff frequency, rejecting signals above that frequency, and a highpass filter passes signals above a cutoff frequency and attenuates signals below it. A band-reject filter suppresses signals within a designed bandwidth passing signals outside the rejection band with minimal loss. A bandpass filter can be described by various performance parameters, including center frequency, passband, passband insertion and return loss, upper stopband, lower stopband, and attenuation within the stopbands.

Transitions from a bandpass filter’s passband to its upper and lower stopbands can be extremely rapid or more gradual, typically described by different filter response types including Butterworth, Chebyshev, and Bessel filters. Each type of bandpass filter exhibits some form of tradeoff. For example, a Butterworth filter is typically characterized by flat amplitude response across its passband, sacrificing sharpness in the transitions from passband to stopbands. Chebyshev filters achieve sharp transitions, at the cost of higher amplitude ripple in the passband than a Butterworth filter. Bessel filters offer linear passband phase response, giving up some stopband attenuation compared to the other two filter types.

The performance of a PCB filter is highly dependent on the circuit substrate material. The choice of material can limit center frequency, passband loss, and other key filter parameters. For many filter designers, the choice of material starts with a laminate’s dielectric constant. For a distributed-element filter such as a microstrip bandpass filter, the size of the transmission lines and distributed filter elements is inversely proportional to the square root of the PCB material’s dielectric constant; in short, PCB materials with higher dielectric constants make it possible to design and fabricate smaller filters for a given frequency. RF/microwave filter designers have long favored PCB substrates with dielectric constants of 10 or higher to create filter circuits with relatively small dimensions for a given center frequency/wavelength.

Since the dimensions of microstrip and other PCB filters are determined by the dielectric constant of a circuit material, it is important that the value of dielectric constant used for a particular material is very accurate. The dielectric constant of any PCB material can vary, so it is critical that these variations remain within the dielectric-constant tolerance range cited for a particular material by its manufacturer, such as 10.2 ± 0.25. Whether a filter’s dimensions are calculated manually or with the help of a computer-aided-design (CAD) program, even small errors in the value of the dielectric constant used in the calculations will result in unwanted changes in designed wavelength/frequency and shifts in center frequency and passband.

Dissipation factor or dielectric loss is another important circuit material parameter for bandpass filters. Quite simply, low values of dissipation factor indicate materials capable of achieving low insertion loss. For a bandpass filter, low PCB dissipation factor also means high filter quality factor (Q), which translates into the potential for a filter with low passband insertion loss and sharper transitions from passband to stopbands.

When designing and fabricating RF/microwave PCB-based bandpass filters, variations in dielectric constant should be minimized whenever possible. A circuit material parameter known as moisture absorption can play a large role in the stability of the material’s dielectric constant under certain environmental conditions, notably under high humidity. Ideally, a PCB material’s moisture absorption should be as low as possible. A material with a high value of moisture absorption can suffer variations in dielectric constant and dissipation factor that far exceed the tolerance ranges specified by the manufacturer. The dielectric constant of the material will change with even a small amount of moisture absorption, resulting in unexpected performance variations in bandpass filter center frequency, passband, and passband insertion loss.

Filter designers choose PCB materials with high dielectric constants in order to minimize the dimensions of their RF/microwave filters. A popular dielectric-constant value for such materials is 10.2, typically for materials based on polytetrafluoroethylene (PTFE). Although a filled PTFE substrate has excellent electrical properties, it can be guilty of moisture absorption on the order of 0.25%. Although this is a relatively small value compared to most PCB materials, a PCB material with this value of moisture absorption can exhibit significant changes in dielectric constant and dissipation factor under conditions of high humidity, possible causing a filter to exceed its performance limits for passband loss or suffer a shift in center frequency and passband from expected values.

RT/duroid 6010.2LM microwave laminate from Rogers Corp. is a composite that blends ceramic filler with PTFE for stable performance and low moisture absorption. The material achieves small bandpass filter dimensions, by merit of its high dielectric constant of 10.2 in the z direction and 10 GHz, with tolerance of ±0.25, and features dissipation factor of only 0.0028 for low passband insertion loss. Its moisture absorption is a fraction of that for many filled PTFE substrates, at typically only 0.01% (compared to 0.25% for other filled PTFE substrates).

A bandpass filter fabricated on this material will have the same dimensions as a filter formed on filled PTFE with dielectric constant of 10.2. However, it will not suffer variations in dielectric constant and dissipation factor, with their resulting variations in filter performance, in environments in which humidity may change dramatically. In fact, the improvements possible with this material compared to PTFE for bandpass filters are detailed in a study available for free download from the Rogers’ web site, “The Benefits of Selecting RT/duroid 6010LM for Band Pass Filter Applications.”

RT/duroid 6010.2LM laminates can be specified with various thicknesses and cladding options and well suited for a wide range of RF/microwave bandpass filters. As the next blog will show, however, other circuit materials with lower dielectric constants and different parameters are available in support of repeatable, high-performance RF/microwave bandpass filters.

Do you have a design or fabrication question? John Coonrod and Joe Davis are available to help. Log in to the Rogers Technology Support Hub and “Ask an Engineer” today.

This post authored by John Coonrod originally appeared on the Rog-Blog hosted by Microwave Journal

Circuit designers select laminates for printed-circuit boards (PCBs) by merit of relative dielectric constant (Dk), among other parameters. Suppliers of laminates furnish Dk values on their data sheets and web sites, but designers often prefer the reassurance of knowing the Dk value as it relates to their specific application.  The last blog explored the way that materials manufacturers typically use four techniques to evaluate the Dk of a dielectric material in its “raw” form, meaning without circuits. This blog will explore some common methods that materials users employ for determining a laminate’s Dk value and focus on a practical method.

Compared to the techniques used by materials manufacturers, RF/microwave circuit designers who use laminates typically rely on fabricating and measuring well-characterized circuits and structures on a material, using circuits and structures with  behavior closely matched to the desired application. These test results can then be compared with the values computed by microstrip design equations, which relate the physical dimensions of transmission lines to electrical parameters, such as frequency and phase.

Some of the more common microwave evaluation circuits include microstrip ring resonators, strip resonators, highly selective filters, and phase delay circuits.  Of course, many options are available and one test method proven to be a good indicator of the performance of a microstrip transmission line is the differential phase-length method.  It is a relatively simple approach and can provide results of Dk performance over a wide range of frequencies.

The microstrip differential phase-length method is based on two transmission-line circuits fabricated on the same material and ideally in close proximity of each other.  The circuits should be identical in every way except physical length. Typically, a long and a short circuit are used, with the difference in length a ratio of 3:1 or greater as a general guideline. The connectors on both circuits should be identical.  Ideally, a pressure contact connector should be used so the same connectors can be used on both circuits. Whenever possible, a low-reflection fixture should be used for the signal launch.

The basic concept for the differential phase-length method is relatively simple.  Two circuits of different physical lengths are measured for their phase responses at a discrete frequency.  The microstrip phase-response formula is then used to calculate the effective dielectric constant (εeff) of the circuit, using the differential length between the two circuits (ΔL) and the differential phase angle (ΔΦ) between the two circuits. The formula can be used to calculate the effective Dk with the given measured data and at that specific frequency:

Once εeff is known, a software routine which also takes into account the circuit geometry can determine the Dk value of the circuit by an iterative process.  When this is determined for one frequency, the process is repeated at the next higher frequency, and so on, ultimately producing a graph of Dk vs. frequency over a wide range of frequencies, as shown in the plot.

10Ml-Laminate-Microstrip-Differential-Phase Graph

A plot of Dk vs. frequency can be a good reference for the Dk value of the material at many different frequencies as well as an indicator as to the dispersive property of the material.  It can be seen in the above graph that this material has very good dispersion. For some materials, the slope of the curve is significantly negative, implying more dispersion; dispersion is a change in Dk relative to a change in frequency.

Each Dk test method has its benefits and shortcoming. The differential phase-length method is a transmission/reflection method which is typically less accurate than resonator methods. However the differential phase-length method provides Dk values over a wide range of frequencies whereas the resonator methods typically yield Dk results at one or more discrete frequencies.

Regardless of the chosen evaluation circuit or Dk test method, the designer should try to best match the type of circuit/method in the test to the type of circuit in their application, to obtain the best approximation of the laminate’s Dk for that application. To assist designers, the next blog will discuss a realistic value of relative dielectric constant known as “design Dk.” It is a value developed for circuit designers, to provide more reliable and accurate results when designing RF/microwave circuits with modern circuit and electromagnetic (EM) software simulation tools.

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